Orthogonal frequency division multiplexing is a well-known technique for transmitting high bit rate digital data signals. Rather than modulate a single carrier with the high speed data, the data is divided into a number of lower data rate channels each of which is transmitted on a separate subcarrier. In this way the effect of multipath fading is mitigated. In an OFDM signal the separate subcarriers are spaced so that they overlap, as shown for subcarriers 12 in spectrum 10 of FIG. 1a. The subcarrier frequencies are chosen that so that the subcarriers are mutually orthogonal, so that the separate signals modulated onto the subcarriers can be recovered at the receiver. One OFDM symbol is defined by a set of symbols, one modulated onto each subcarrier (and therefore corresponds to a plurality of data bits). The subcarriers are orthogonal if they are spaced apart in frequency by an interval of 1/T, where T is the OFDM symbol period.
An OFDM symbol can be obtained by performing an inverse fourier transform, preferably an Inverse Fast Fourier Transform (IFFT), on a set of input symbols. The input symbols can be recovered by performing a fourier transform, preferably a fast fourier transform (FFT), on the OFDM symbol. The FFT effectively multiplies the OFDM symbol by each subcarrier and integrates over the symbol period T. It can be seen that for a given subcarrier only one subcarrier from the OFDM symbol is extracted by this procedure, as the overlap with the other subcarriers of the OFDM symbol will average to zero over the integration period T.
Often the subcarriers are modulated by QAM (Quadrature Amplitude Modulation) symbols, but other forms of modulation such as Phase Shift Keying (PSK) or Pulse Amplitude Modulation (PAM) can also be used. To reduce the effects of multipath OFDM symbols are normally extended by a guard period at the start of each symbol. Provided that the relatively delay of two multipath components is smaller than this guard time interval there is no inter-symbol interference (ISI), at least to a first approximation.
FIG. 1b shows an exemplary OFDM transmitter 100 (here in a mobile terminal, MT) and an exemplary OFDM receiver 150 (here in an access point, AP). In the transmitter 100 a source 102 provides data to a baseband mapping unit 104, which optionally provides forward error correction coding and interleaving, and which outputs modulated symbols such as QAM symbols. The modulated symbols are provided to a multiplexer 108 which combines them with pilot symbols from a pilot symbol generator 106, which provides reference amplitudes and phases for frequency synchronisation and coherent detection in the receiver (in other arrangements differential detection may be employed). The combination of blocks 110 converts the serial data stream from multiplexer 108 to a plurality of parallel, reduced data rate streams, performs an IFFT on these data streams to provide an OFDM symbol, and then converts the multiple subcarriers of this OFDM symbol to a single serial data stream. This serial (digital) data stream is then converted to an analogue time-domain signal by digital-to-analogue converter 112, up-converted by up-converter 114, and after filtering and amplification (not shown) output from an antenna 116. Antenna 116 may comprise an omni-directional antenna, a sectorised antenna or an array antenna with beamforming.
The signal from antenna 116 of transmitter 100 is received by an antenna 152 of receiver 150 via a “channel” 118. Typically the signal arrives at antenna 152 as a plurality of multipath components, with a plurality of different amplitudes and phases, which have propagated via a plurality of different channels or paths. These multipath components combine at the receiver and interfere with one another to provide an overall channel characteristic typically having a number of deep nulls, rather like a comb, which generally change with time (particularly where the transmitter or receiver is moving). Often there will be a number of transmitters in the same general location, for example an office, and this gives rise to co-channel interference, which can be more problematic than multipath.
The antenna 152 of receiver 150 is coupled to a down-converter 154 and to an analogue-to-digital converter 156. Blocks 158 then perform a serial-to-parallel conversion, FFT, and parallel-to-serial re-conversion, providing an output to demultiplexer 160, which separates the pilot symbol signal 162 from the data symbols. The data symbols then demodulated and de-mapped by base-band de-mapping unit 164 to provide a detected data output 166. Broadly speaking the receiver 150 is a mirror image of the transmitter 100. The transmitter and receiver may be combined to form an OFDM transceiver.
OFDM techniques may be employed in a variety of applications and are used, for example, for military communication systems and high definition tv. Here, applications of the invention will be discussed with particular reference to the HIPERLAN (High Performance Radio Local Area Network) Type 2 standard (www.etsi.org/technicalactiv/hiperlan2.htm, and DTS/BRAN-0023003 v 0.k). Although applications of the invention are not limited to this environment HIPERLAN 2 wireless local area network communications are managed by a common node, the access point.
The receiver of FIG. 1b is somewhat simplified as, in practice, there is a need to synchronise the FFT window to each OFDM symbol in turn, to avoid introducing non-orthogonality and hence Inter-Carrier Interference (ICI). This may be done by auto-correlating an OFDM symbol with the cyclic extension of the symbol in the guard period but it is generally preferable, particularly for packet data transmission, to use known OFDM (training) symbols which the receiver can accurately identify and locate, for example using a matched filter. It will be appreciated that this matched filter operates in the time domain, that is before the FFT is carried out (as opposed to the post-FFT frequency domain). In a packet data system data packets may be provided with a preamble including one or more of these training symbols.
FIGS. 2a and 2b show, respectively, a receiver front end 200 and receiver signal processing blocks 250 of a HIPERLAN 2 mobile terminal (MT) OFDM receiver. The receiver 250 shows some details of the analogue-to-digital conversion circuitry 252, the synchronisation, channel estimation and control circuitry 254 and the de-packetising, de-interleaving and error correcting circuitry 256.
The front end 200 comprises a receive antenna 202 coupled to an input amplifier 204 and a mixer 206, which has a second input from an IF oscillator 208 to mix the RF signal to IF. The IF signal is then provided to an automatic Automatic Gain Control (AGC) amplifier 212 via a band pass filter 210, the AGC stage being controlled by a line 226 from control circuitry 254, to optimise later signal quantisation. The output of AGC 212 provides an input to two mixers 214, 216, which are also provided with quadrature signals from an oscillator 220 and splitter 218 to generate quadrature I and Q signals 222, 224. These I and Q signals are then over-sampled, filtered and decimated by analogue-to-digital circuitry 254. The over-sampling of the signal aids the digital filtering, after which the signal is rate reduced to the desired sample rate.
It is desirable (but not absolutely essential) to compensate for the effects of the transmission channel. This can be done using a known symbol, for example in preamble data or one or more pilot signals. In the receiver 250 of FIG. 2 a known preamble symbol, referred to as the “C symbol”, is used to determine a channel estimate. The receiver synchronises to the received signal and switch 258 is operated to pass the received C symbol to channel estimator 260. This estimates the effect of the channel (rotation of the symbols in the sub-carriers) on the known C symbol so that the effects of the channel can be compensated for, by multiplying by the complex conjugate of the channel response. Alternatively the one or more pilot signals (which also contain known symbols) can be used to determine a channel estimate. Again the phase rotation and amplitude change required to transform the received pilot into the expected symbol can be determined and applied to other received symbols. Where more than one pilot is available at more than one frequency improved channel compensation estimates can be obtained by interpolation/extrapolation to other frequencies using the different frequency pilot signals.
In FIG. 2 the receiver front end 200 will generally be implemented in hardware whilst the receiver processing section 250 will often be implemented in “software”, as illustrated schematically by Flash RAM 262 using, for example, ASICs, FPGAs or one or more DSP (digital signal processor) chips. A similar division between hardware and software will generally be present in the transmitter. However the skilled person will recognise that all the functions of the receiver of FIG. 2 (or of an equivalent transmitter) could be performed in hardware. Similarly the exact point at which the signal is digitised in a software radio will generally depend upon a cost/complexity/power consumption trade-off, as well as upon the availability of suitable high speed analogue/digital converters and processors, and that the RF signal could be digitised at IF or a higher frequency.
FIG. 3 shows an example of a Media Access Control (MAC) frame 300 of a packet data communications system including preamble sequences. The MAC frame includes a broadcast channel (BCH) burst 302, a frame channel (FCH) burst 304, an access feedback channel (ACH) burst 306, a down-link (DL) burst 308, an up-link (UL) burst 310, a direct link (DiL) burst 312, and a random access (RCH) burst 314, all of which contain a preamble sequence.
FIGS. 4a to e show, respectively, a broadcast burst, downlink burst, an uplink burst with a short preamble, uplink burst with a long preamble, and a direct link burst of a HIPERLAN 2 physical layer signal. Each of these bursts comprises a preamble portion 400 and a data payload portion 402. The preamble portions 400 comprise one or more of three basic OFDM symbols, denoted A, B and C. The values of these symbols are known and A and B (and, if desired, C) can be recovered in the time domain (pre-FFT). These symbols are generally used to establish the frame and frequency synchronisation and to set the FFT window for the data following the symbols; they may also be employed to control AGC stage 212. In the receiver of FIGS. 2 A and B are recovered in the time domain and C is recovered in the frequency domain, that is post-FFT.
FIG. 5 illustrates, schematically, the use of these (known) preamble symbols for frame detection 502 based on RSSI (Received Signal Strength Indication), automatic gain control 504, frame synchronisation 506, and frequency synchronisation 508; a schematic illustration of the preamble portion of an MAC frame 500 is also illustrated for comparison.
FIG. 6 shows a plot 600 in the frequency and time domain illustrating the relative positions of preamble sequences 602, pilot signals 604, and data signals 606 for HIPERLAN 2, which has 48 data sub-carriers and 4 pilots (and one unused, central carrier channel 608). As can be seen from FIG. 6 the first four OFDM symbols comprise preamble data, and the pilot signals 604 continue to carry their preamble symbols. However on the remaining (data-bearing) sub-carriers OFDM symbols 5 onwards carry data. In other OFDM schemes similar plots can be drawn, although the preamble and pilot positions may vary (for example, the pilots need not necessarily comprise continuous signals).
It has previously been mentioned that OFDM is a useful technique for alleviating the effects of frequency selective fading caused by multipaths. However with particularly high data rates or in particularly severe multipath environments OFDM communications systems can still suffer from the effects of multipath fading. Moreover in indoor wireless environments, such as small office wireless LANs, there will often be a number of similar systems operating simultaneously in the same frequency band, because of limited spectrum availability. This can result in severe co-channel interference.
One technique which has been proposed for combatting such multipath and co-channel interference is the use of a sectorised transmit and/or receive antenna. The region to be covered is divided into a number of sectors, typically 3, 4 or 6, and one antenna (or more where diversity is employed) is provided for each sector, the patterns of the antennas being arranged to each cover mainly just one sector. In effect the main beam of each of the sector antennas points in a different direction and by selecting the transmit and/or receive direction the effects of multipath components and/or co-channel interference arriving from unwanted directions can be reduced. HIPERLAN 2, for example, supports the use of up to seven sectors at the Access Point. Some of the benefits of employing a sectorised switching array antenna in a HIPERLAN 2 environment are described in “Performance of HIPERLAN 2 using Sectorised Antennas” A. Dufexi, S. Armour, A. Nix, P. Karlsson and D. Bull, IEE Electronics Letters Feb. 15, 2001, volume 37 no. 4, page 245.
Another approach employed to mitigate the effects of multipath and co-channel interference uses a beamforming antenna array, such as a linear array of antenna elements in which the inter-antenna spacing is approximately one half a (carrier) wavelength. Signals from the antennas are combined, with appropriate phase and amplitude weightings, to provide a combined response with one or more lobes or beams. An array comprising n elements can be arranged to provide up to n−1 beams.
There are a number of different beamforming algorithms which may be applied to such an adaptive antenna array and details of these will be well known to the skilled person. One commonly used algorithm is the Constant Modulus Algorithm (CMA), described in J. R. Treichler and B. G. Agee, “A New Approach to Multipath Correction of Constant Modulus Signals”, IEEE Trans. Acoust. Speech and Signal Process., vol. ASSP-31, no. 2, page 459, 1983, which is hereby incorporated by reference. Broadly speaking this algorithm iteratively determines the weights for combining the signals from the antenna elements based upon a cost function chosen to make the spectrum of the combined signals approximately flat. The phase angles of the weights are chosen so that the beams point in the direction of maximum signal power, or, alternatively, so that nulls are formed in the directions of the unwanted multipath components or co-channel interference.
Determining appropriate weights for the antenna array elements is relatively straightforward in a narrow band system but in an OFDM receiver, where the bandwidth occupied by the group of sub-carriers is normally >1 MHz and in many cases >10 MHz, a single set of weights is unlikely to be optimal across the entire bandwidth and may only be valid, for example, at the centre of the frequency band. This can be understood, for example, from the consideration that the antenna element spacing, in terms of fractions of a sub-carrier wavelength, varies across the OFDM frequency band. In the receiver of FIG. 1 adaptive array weights may be applied at points 168, 170, or 172 but applying the array weights at positions 168 or 170 (pre-FFT) will not normally result in a good set of estimated weights across the frequency band.
One solution to this problem is therefore to apply weightings after the FFT, at point 172, where different sets of weights can be applied to each sub-carrier. FIG. 7 shows an OFDM receiver 700 in which a separate set of weights is applied to each sub-carrier in this way. However it will be appreciated that with K sub-carriers and L antenna elements a total of K×L weights must be determined, which is a lengthy and processor-intensive task adding considerably to the receiver complexity. EP 0 852 407 describes an arrangement in which an operational band is partitioned into four equal sub-bands, one set of weights being calculated for each sub-band rather than for each sub-carrier, to reduce the number of weights to be calculated. However this is still a relatively complicated procedure and, moreover, produces a sub-optimal result. An alternative approach is described in Fujimoto et al, “A Novel Adaptive Array Utilising Frequency Characteristics”, IEICE Trans. Commun., vol. E 83-B, no. 2 Feb. 2000, page 371, which is hereby incorporated by reference, in which the post-FFT separated sub-carriers are used to determine a single set of pre-FFT time domain weights using CMA. This approach provides a considerable simplification of the weight determining procedure but, again, the weights are sub-optimal.
U.S. Pat. No. 6,249,250 describes OFDM adaptive antenna weight determination techniques for use in either the time or frequency domain (but not both simultaneously). Other weight-determination techniques involving sub-carrier clustering are described in Japanese patent application number 2000-391221 filed on 22 Dec. 2000, inventor Hidehiro Matsuoka, and British patent application number 0108026.6 filed on 30 Mar. 2001 entitled “Adaptive Antenna”.
The above-described weight calculation techniques for array antennas each have pros and cons, some providing relatively accurate weight determination at the expense of complex and time-consuming processing, others having simpler, faster weight determination algorithms but in general providing poorer weight estimates. There is therefore a need for improved array antennas and weight determination techniques which can provide good weight estimates without imposing an excessive signal processing burden.